Transmitter Signal Shaping

ABSTRACT

A transmitter comprises at least one nonlinear circuit, and a power spectral density (PSD) shaping circuit. The PSD shaping circuit is operable to receive a symbol of a modulated signal, wherein the symbol corresponds to a first one or more frequency bins. The PSD shaping circuit is operable to perform iterative processing of the symbol, wherein each iteration of the processing comprises: generation of a pre-distortion signal based on a model of the at least one nonlinear circuit, wherein the nonlinearly distorted signal corresponds to a second one or more frequency bins; and combination of the symbol, or a pre-distorted version of the symbol, with the nonlinearly-distorted signal. The generation of the pre-distorted signal may comprise generation of a nonlinearly-distorted signal, and adjustment of one or more components of the nonlinearly-distorted signal.

PRIORITY CLAIM

This application is a continuation of U.S. application Ser. No.14/687,861 filed Apr. 15, 2015 (now U.S. Pat. No. 9,246,523) whichclaims priority to U.S. provisional patent application 62/042,356 filedon Aug. 27, 2014, and U.S. provisional patent application 62/126,881filed on Mar. 2, 2015. Each of the above mentioned documents is herebyincorporated herein by reference in its entirety.

BACKGROUND

Limitations and disadvantages of conventional and traditional approachesto electronic communications will become apparent to one of skill in theart, through comparison of such systems with some aspects of the presentinvention as set forth in the remainder of the present application withreference to the drawings.

BRIEF SUMMARY

Systems and methods are provided for transmitter signal shaping,substantially as shown in and/or described in connection with at leastone of the figures, as set forth more completely in the claims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 is a diagram illustrating an example transmitter with TXnonlinear shaping circuit.

FIG. 2 is a diagram illustrating additional details of an exampletransmitter with Tx nonlinear shaping circuit.

FIGS. 3A-3D depict example transmitters which uses a PA PSD shapercircuitry with or without other aspects of the TX nonlinear shapingcircuitry of FIG. 1.

FIG. 4 illustrates an example response of digital nonlinear functioncircuit.

FIG. 5 is a diagram illustrating control of the PA PSD Shaper inaccordance with an example implementation of this disclosure.

FIG. 6A illustrates an alternate implementation of the transmitter ofFIG. 3A.

FIG. 6B illustrates an alternate implementation of the transmitter ofFIG. 3B.

DETAILED DESCRIPTION

Conventional receivers can manage relatively low amounts of nonlineardistortion of their input communication signal. In contrast, receiversin accordance with various implementations of this disclosure may becapable of handling deep communication signal distortion by measuringand modeling the nonlinear response. This capability allowscommunication systems to handle deep Power Amplifier (PA) nonlinearity.This capability can also be used to handle intentional distortiondigitally applied by the transmitter in order to improve various signalcharacteristics. Circuitry for introducing such digital distortion isreferred to herein as a TX nonlinear shaper. In various implementations,the TX nonlinear shaper can achieve one or more the following goals:

-   -   Reduce signal peak to average power ratio, which has many uses.        One use applicable to working with deep PA distortion and        unraveling it at the receiver using PA model is preventing the        signal from getting too close to the power rails. An advantage        of this is that PA modeling near the power rails may be        difficult (e.g., costly in terms of time, memory, processing        power, etc.).    -   Reduce signal cubic metric (a metric defined by 3GPP) in order        to reduce transmitted signal out-of-band distortion (i.e. to        reduce occurrence of exceeding allowed out-of-band spectral        emission mask or Adjacent Channel Leakage Ratio), while working        relatively close to PA compression.    -   Allowing to work at lower PA output backoff for a particular        level of receiver performance    -   Trading off, at the receiver, noise enhancement due to a        distortion cancellation process in the receiver, vs Minimizing        received signal clips due PA signal exceeding PA rails.    -   Optimizing distortion function for operation with a receiver        that is capable of handling nonlinear distortion. Where the        optimizing may, for example, be in the sense of maximum coded        signal transmission performance.

The composite response of the TX nonlinear shaper and the PA is referredto in this disclosure as the composite nonlinear response. Thiscomposite nonlinear response may include memory. As compared to DigitalPre-distortion (DPD) techniques which attempts to substantiallylinearize the PA response, thus producing a linearized compositeresponse (typically referred to as “soft limiter”), the TX nonlinearshaper of various implementations of this disclosure aims for anonlinear composite response that is soft (well behaved derivative) andmonotonic, and allows substantially better receiver performance (for areceiver that can manage distorted signal) than the linearized compositeachieved with conventional DPD.

Referring now to FIGS. 1 and 2, in an example implementation, the uplink(UL) transmitter (e.g., of each of multiple mobile units) have thegeneral block diagram shown in FIGS. 1 and 2. It is understood that thesame principles described by this disclosure can be applied in otherorthogonal frequency division multiplexed (OFDM) or frequency divisionmultiple access (FDMA) systems such as orthogonal frequency divisionmultiple access (OFDMA), and single carrier FDMA (LTE ScFDMA). Theexample transmitter 101 comprising forward error correction (FEC)encoding circuit 110, interleaving circuit 112, constellation mappingcircuit 114, bin mapping circuitry 116, buffer 118, inverse discreteFourier transform (IDFT) circuit 120, upsampler 122, Tx nonlinear shapercircuit 100, parallel to serial conversion (P/S) and cyclic prefix (CP)insertion circuit 124, digital filtering and/or windowing circuitry 128,digital-to-analog conversion (DAC) and lowpass filter circuit 128, andpower amplifier (PA) circuit 130.

The FEC encoding circuit 110 is operable to encode an incoming bitstreamin accordance with any suitable FEC encoding algorithm such asReed-Solomon encoding, low density parity check (LDPC) encoding, turboencoding, and/or the like to output FEC codewords.

The interleaving circuit 112 is operable to interleave the FECsymbols/bits output by FEC encoding circuit 110 such that the order inwhich the FEC symbols/bits emerge from the interleaving circuit 112 isdifferent than the order in which they emerge from the FEC encodingcircuit 110.

The constellation mapping circuit 114 is operable to map bits of the FECcodewords to symbols of a selected constellation (e.g., BPSK, QPSK,N-QAM, or the like).

The bin mapping circuitry 116 is operable to assign each of thefrequency-domain symbols output by the constellation mapping circuit 114to a respective one of a plurality of frequency bins. The frequencymapping may, for example, be based on frequency bins assigned to thetransmitter 101 by a link partner. For example, where the transmitter101 is in a LTE handset, the frequencies may be assigned by the basestation currently handling the handset. The output of the bin mappingcircuitry 116 may be zero padded relative to its input.

The buffer 118 is operable to store the frequency-mapped symbols untilan OFDM symbol's worth of constellation symbols (e.g., an integer numberF) are buffered, at which point the constellation symbols are passed tothe IDFT circuit 120.

The IDFT circuit 120 converts the frequency-domain representation of theconstellation symbols to a time-domain representation (e.g., comprisingT time-domain samples).

The upsampler 122 is operable to upsample the time-domain representationby a factor of L (e.g., resulting if L×T samples from the original Tsamples).

The P/S and CP add circuit 124 is operable to convert the parallelstream of time-domain samples to a serial stream and then adds a cyclicprefix.

The digital filtering and/or windowing circuit 126 is operable to applya windowing function and/or other filtering to the output of circuit124, resulting in a digital-domain OFDM symbol.

The DAC and lowpass filtering circuit is operable to filter and convertto analog the digital-domain OFDM symbol output by circuit 126,resulting in an analog-domain OFDM symbol.

The power amplifier (PA) 130 is operable to transmit the OFDM symbolonto a wired, wireless, or optical communication medium.

In the example implementation shown in FIGS. 1 and 2, the Tx nonlinearshaper circuit 100 comprises digital nonlinear function (DNF) circuit102, DNF power spectral density (PSD) shaper 104, PA compensationcircuit 106, and iterative PA PSD shaper circuit 108. Each of thesecircuits is described in further detail below.

In the example transmitter of FIGS. 1 and 2, the input bit stream isencoded by FEC encoder 110, interleaved by interleaver 112, mapped tosymbols according to a selected constellations (i.e. QAM mapping) bymapping constellation mapping circuit 114, and then mapped in frequencyby frequency mapping circuit 118 to subcarriers allocated to thetransmitter 101. These subcarriers are typically allocated by the basestation, and may or may not be continuous in frequency. Inverse discreteFourier transform (IDFT) circuit 120 operates to transform the signalfrom frequency domain to time domain and the signal is then interpolatedby interpolator 122. The resulting oversampled signal x(n) is thenpassed through the digital nonlinear function (DNF) circuit 102, inorder to reduce its PAPR and/or Cubic Metric and improve nonlinearreceiver performance. The resulting digitally compressed signal y(n) isfed to the DNF PSD shaper 104 that substantially cancels the out-of-bandre-growth due to distortion introduced by DNF circuit 102. The signalz(n) output by PSD shaper circuit 104 is processed by PA compensationcircuit 106 which uses a gross model of the PA 130 to approximatelycompensate for the nonlinearity of the PA 130. The effect of the PAcompensation circuitry may be a composite response of the PAcompensation circuit 106 and PA 130 that is a soft limiter (i.e.,exhibits a clipped linear response). This composite may not need to bean accurate soft limiter, since it just serves to transform thecomposite response of the DNF 102, the PA compensation circuit 106, andthe power amplifier (PA) such that it approximates the nonlinearresponse targeted by the DNF circuit 102. Therefore the PA compensationcircuit 106 can be far less accurate than conventional digitalpre-distortion (DPD) functions. The PA compensated signal s(n) is fed tothe iterative PA PSD shaper 108, which may operate, when an accuratemodel of the PA 130 is known, to cancel out the out-of-band distortionproduced by the PA 130. A cyclic prefix (CP) is then added by circuit124 to the output of the iterative PA shaper 108, and windowing may beapplied by circuit 126. The resulting signal is DAC converted andfiltered by circuit 128, up-converted and amplified by the PA 130.

I. DNF Circuit 102

The DNF circuit 102 may be optimized to compress the transmitted signalto limit out-of-band distortion introduced by the Power Amplifier (e.g.reducing cubic metric) and maximize receiver performance. In variousexample implementations, the DNF circuit 102 results in a smooth andmonotonic nonlinearity designed to allow operation under deepcompression at the Power Amplifier 130 without violating transmissionmask and still allowing for sufficient receiver performance. A fewexample Approaches to configuring the DNF circuit 102 will now bediscussed. It will be understood, however, that many design approachesare possible and fall within the scope of this disclosure.

The following is notation used in this disclosure.

-   -   1. x(n)—Original transmission signal    -   2. y(n)—Distorted signal    -   3. g_(NL)—Digital Nonlinear function (DNF)    -   4. ƒ_(NL)—PA nonlinearity    -   5. c_(NL)—Composite nonlinearity    -   6. X_(sat)—Lowest x value that is saturated by none linearity,        i.e.

c _(NL)(X)=c _(NL)(x _(SAT))=A _(SAT) for any |x|≧x _(SAT)

-   -   7. h_(NL)—inverse function    -   8. A_(SAT)—the maximum PA output signal.    -   9. CM—Is the desired cubic metric of the transmission signal.    -   10. B_(PA) _(_) ₀—The PA Output Backoff    -   11. σ_(x)—RMS of input OFDM signal to DNF    -   12. v′,σ_(n)—Receiver input noise and its standard deviation of        receiver input noise    -   13. v,σ_(n)—Nonlinear-Solver Receiver output noise and its        standard deviation of receiver enhanced noise due to receiver        distortion cancellation    -   14. σ_(vd)—Standard deviation of total distortion and noise at        output of distortion cancelling receiver

A. Example 1

In a first example implementation, the DNF 102 aims to minimize theaverage total noise and clip distortion experienced by the receiver inthe process of cancelling out the distortion. In such an implementation,the DNF 102 approximately models distortion cancellation at the receiveras an inversion of the composite c_(NL). In other words, the receiveddistorted signal is

Y _([n]) =c _(NL)(x _([x]))

Thus, the inverting function is denoted h_(NL) (y) such that

h _(NL)(y _([n]))=h _(NL)(c _(NL)(x _([n]))+v′ _([n]))≈′x _([n])

Based on AM/AM and AM/PM distortion curves for the inversion of thecomposite c_(NL), the amplitude of the received samples can be used,i.e.

h _(NL)(y _([n]))=h _(amp)(|Y[n]|)·h _(phase)(|y[n]|)·exp(j·arg(y))

where h_(amp) and h_(phase) compute the correct amplitude and phasecorrection.

The inversion, however, is noisy and biased, two impairments may bedesirable to minimize

h _(NL)(y _([n]))=h _(NL)(c _(NL)(x _([n]))+v′ _([n]))=x _([n]) +v_([n]) +b _([n])

where b_([n]) is the bias occurring when |x_([n])|>x_(sAT), since inthis case y_([n]) cannot be inverted since x_([n]) is saturated byc_(NL)(x).

To estimate the noise enhancement, the basic identity

${h^{\prime}( {c_{NL}(x)} )} = \frac{1}{c_{NL}^{\prime}(x)}$

can be used, which can be easily derived by differentiatingh(c_(NL)(x))=x. Where ƒ′( ) denotes the derivative of function ƒ( ).

Thus the total noise+distortion due to clipping is

$\sigma_{vd}^{2} = {{\frac{2}{\sigma_{x}^{2}}{\overset{x_{SAT}}{\int\limits_{0}}{{x \cdot {\exp ( {- \frac{x^{2}}{\sigma_{x}^{2}}} )} \cdot \frac{\sigma_{n}^{2}}{( {c_{NL}^{\prime}(x)} )^{2}}}{x}}}} + {\frac{2}{\sigma_{x}^{2}}{\overset{\infty}{\int\limits_{X_{SAT}}}{{x \cdot {\exp ( {- \frac{x^{2}}{\sigma_{x}^{2}}} )}}( {x - x_{SAT}} )^{2}{x}}}}}$

Since the inversion is based on the received signal amplitude, and theOFDM signal is complex Gaussian, the underlying distribution determiningthe noise enhancement is Rayleigh (with corresponding complex Gaussiansignal power of σ_(X) ²).

An example process for configuring the DNF circuit 102 for minimizingaverage total noise and distortion is as follows:

Using Lagrange multipliers it can be shown that a good choice for thecomposite function c_(NL)(x) would be such that:

${c_{NL}^{\prime}(x)}^{3} = {{{K \cdot x \cdot {\exp ( {- \frac{x^{2}}{2 \cdot \sigma_{x}^{2}}} )}}\mspace{14mu} {for}{\mspace{11mu} \;}x} < X_{SAT}^{\prime}}$

And to bound |c_(NL)(x)|A_(SAT) K is determined such that

$A_{SAT} = {\overset{X_{SAT}^{\prime}}{\int\limits_{0}}{{c_{NL}^{\prime}(x)}{x}}}$

The parameter X′_(SAT) may be optimized numerically by searching for theX′_(SAT) that minimizes σ_(vd) ²(X′_(SAT))

B. Example 2

In another example implementation using a more heuristic criteria, theDNF circuit 102 is used in order transform the complex Gaussiandistribution of x_([n]) at the input of the DNF circuit 102 to anotherdistribution having preferred characteristics at the output of the PA130. For example, the output distribution may be configured to maximizecapacity (or achieve a desired minimum threshold capacity) for theparticular power backoff at which the PA 130 is operating. For example,for very low PA output backoff of 3 dB and high signal-to-noise ratio(SNR) at the receiver, the system could transform the input distributionsuch that the distribution at the output of PA 130 is uniform within acircle of a radii A_(SAT). This maximizes the entropy of thepower-limited transmitted signal, which approximately maximizes capacityin the assumed high SNR case.

The distribution of y=C_(NL)(x) when x is complex Gaussian in againinfluenced by the derivative c′_(NL)(x). Using the basic identity p_(y)(y)=p_(x)(c_(NL) ⁻¹(y))/c_(NL)′(c_(NL) ⁻¹(y)), then to get a uniformdistribution the following is needed:

$\begin{matrix}{1 = {p_{y}(y)}} \\{= {\frac{1}{{\pi\sigma}_{x}^{2}}{\exp ( {- \frac{{c_{NL}^{- 1}(y)}^{2}}{\sigma_{x}^{2}}} )}\frac{1}{c_{NL}^{\prime}( {c_{NL}^{- 1}(y)} )}}}\end{matrix}$

Thus to output a uniform distribution the derivative c′_(NL)(x) is setas follows:

${c_{NL}^{\prime}(x)} = {K \cdot {\exp ( {- \frac{{c_{NL}^{- 1}(y)}^{2}}{\sigma_{x}^{2}}} )}}$

and to bound |c_(NL)(x)|≦A_(SAT) it is also needed to determine K suchthat

$A_{SAT} = {\overset{X_{SAT}^{\prime}}{\int\limits_{0}}{{c_{NL}^{\prime}(x)}{x}}}$

As before, the system can select X_(SAT)′ to minimize the overall clipdistortion, or optimize it based on receiver performance.

C. Example 3

The approaches in examples 1 and 2 above have a general form ofc_(NL)′(x)=K·pdƒ(x), where the probability density function (pdƒ(x)) isthe underlying Rayleigh or complex Gaussian probability densityfunction. These approaches can be generalized to:

c _(NL)′(x)=K·pdƒ(x)^(α)

where α is a real positive value, pdƒ(x) is some probability densityfunction, and K is chosen such that

$A_{SAT} = {\overset{X_{SAT}^{\prime}}{\int\limits_{0}}{{c_{NL}^{\prime}(x)}{x}}}$

This family of DNF functions contains example 2 (above) when pdƒ(x) isGaussian and α=1. Empirically, the whole set of functions gives a goodtradeoff between noise enhancement and backoff.

As can be seen in FIG. 4, the example DNF functions (for α=0.5 andα=0.2) have larger slope at X=0 than a conventional soft limiter,therefore providing lower backoff and cubic metric. On the other hand,they have soft behavior near A_(SAT), thus improving receiverperformance. Empirically the soft limiter performance achieved byconventional DPD is lower than the performance achieved using themethods and systems described herein.

In example implementations 1-3, above, the composite c_(NL)( ) has beenoptimized. The composite, however, accounts for the responses of boththe DNF 102 and the PA 130. Thus, it is needed to compensate DNF for PAexistence to get the desired composite. In other words,ƒ_(NL)(g_(NL)(x))=c_(NL)(x)=>g_(NL)(x)=ƒ_(NL) ⁻¹(c_(NL)(x)), whereg_(NL)(x) is nonlinear response of the DNF circuit 102. Accuratemodeling of the PA 130 is not needed for this purpose. One alternativeis to get desired properties of the DNF circuit 102 itself. For example,with low input backoff to the PA 130, the system could optimizeg_(NL)(x) directly instead of c_(NL)( ).

D. Example 4

In another example implementation, criteria of optimizing g_(NL)( ) maybe used to allow mask compliance at higher transmit power. In someinstances, such an optimized g_(NL)( )(e.g., optimized using numericalmethods) outperforms conventional DPD in terms of mask compliance.

E. Example 5

In another example implementation, g_(NL)( ) may be optimized (e.g.,using numerical methods) to relax the requirements of PA PSD Shaper 108.For example, by optimizing g_(NL)( ) to reduce distortion at highfrequency offsets, the requirements of the PA PSD shaper 108 at highfrequency offset are relaxed and therefore the PA PSD Shaper 108 mayoperate at lower overall sampling rate.

II. DNF PSD Shaper Circuit 104

The DNF PSD shaper circuit 104 may operate to reject out-of-banddistortion introduced by the DNF 102. In FIGS. 1 and 2, the DigitalNonlinear Function (DNF) PSD shaper 104 is located right after the DNFcircuit 102 and is used to reject distortion components generated by theDNF circuit 102. Since the DNF circuit 102 operates digitally, thedistortion component it generates are known exactly and therefore can becompletely cancelled before being input to the DAC. Some exampleimplementations of the DNF PSD shaper 104 are discussed next.

In a first example implementation, the DNF PSD shaper 104 is operable tocompute the distortion in the frequency domain and cancel it. This maycomprise the DNF PSD shaper 104 performing discrete Fourier transform(DFT) on the oversampled DNF output y(n) (e.g. at 2× oversampling) toobtain high frequency content of the output of the DNF circuit 102,which contains out-of-band (OOB) components introduced by the DNFcircuit 102. The DNF PSD Shaper 104 may then zero, or attenuate below anapplicable spectral mask, the OOB components. The DNF PSD shaper 104 maythen perform an inverse DFT (IDFT) to convert the signal back to thetime domain.

One drawback with the example implementation just discussed, is thatcancelling the OOB components increases the PAPR/cubic metric of thesignal. Accordingly, in a second example implementation, the DNF PSDShaper 104 adjusts the filtering/zeroing of the OOB components based onthe resulting cubic metric (e.g., attempts to keep the increase in cubicmetric within determined bounds).

In another example implementation, the DNF PSD shaper 104 may activelygenerate an OOB cancelling signal d_(ic)[x] that, on one hand,attenuates the OOB components, and on the other hand does not increasePAPR/Cubic Metric. This can be expressed as follows:

CM(y[n]+d _(ic) [n])≦CM ₀

and

|F(y[n]+d _(ic) [n])−F(x[n])|²≦DistortionMask(ƒ)

where CM( ) computes the cubic metric; CM₀ is a target cubic metric(e.g. 1.3 dB); and DistortionMask(ƒ) is the maximum allowed DNFdistortion mask. For in-band frequencies (as defined by the applicablestandard, and which may include some guard band), DistortionMask(ƒ)takes into account the EVM and in-band emissions. For out-of-bandfrequencies, DistortionMask(ƒ) takes into account adjacent channelleakage ratio (ACLR), and spectral emissions requirements, and alsoincorporates some margin to allow for distortion introduced by the PA130.

III. PA Compensation Circuit 106

The foregoing discussed ways to optimize the DNF circuit 102 such thatsignal characteristics at the output of the PA are improved. Since thePA 130 is part of the composite response from the input of the DNFcircuit 102 to the output of the PA 130, in some implementations it isdesirable or necessary for the transmitter 101 to compensate for the PA130, in order to get the desired composite response. As compared toconventional DPD, which turns the composite response into a soft limiter(linear response that is clipped at some level), the PA compensationcircuit 106 is not a soft limiter, and can use a much less accuratemodel of the PA 130.

An aim of the PA compensation circuit 106 can be expressed as

ƒ_(NL)(g _(NL)(x))=c _(NL)(x)

Several example implementations of the PA compensation circuit 106 willnow be described.

A first example implementation, already discussed above, directlyincorporates the response of the PA compensation circuit 106 into theconfiguration/operation of the DNF circuit 102. By settingg_(NL)(x)=ƒ_(NL) ⁻¹(c_(NL)(x)), then c_(NL)(x)=ƒ_(NL)(g_(NL)(x)). Withthis approach, the transmitter 101 doesn't need a separate PAcompensation circuit 106 in order to comply with an applicable mask andenable sufficient receiver performance.

In another example implementation, the PA compensation circuit 106 isseparated from the DNF circuit 102 and the PA Compensation circuit 106is applied to the output of the DNF PSD Shaper 104. In other words, theprocessing performed by the PA compensation circuit 106 iss_([n])=ƒ_(NL) ⁻¹(z_([n])) (where ƒ_(NL)( ) may be an approximation). Anadvantage of the PA compensation circuit 106 operating on the output ofthe DNF PSD Shaper 104 is that is reduces OOB distortion introduced bythe PA 130.

IV. Iterative PA PSD Shaper Circuit 108

FIG. 2 shows details of an example implementation of the PA PSD shaper108. The example implementation comprises a discrete Fourier transform(DFT) circuit 208, an inverse DFT (IDFT) 210, a composite nonlineardistortion modeling circuit 212, DFT circuit 214, a bin modificationcircuit 216, a memory 218, a switch 220, and a combiner 222.

The DFT circuit 208 is operable to convert the time-domain signal s(n)to frequency-domain signal S[k] (e.g., using a fast Fourier transformalgorithm). The IDFT circuit 210 is operable to convert thefrequency-domain signal A[k] to time-domain signal a(n). The discreteFourier transform (DFT) circuit 208, an inverse DFT (IDFT) 210 may besynchronized to OFDMA or single carrier frequency division multipleaccess (ScFDMA) symbol time, such that out-of-band corrections would beorthogonal to the desired signal.

The composite nonlinear distortion modeling circuit 212 processes thetime-domain signal a(n) to distort the signal using the compositenonlinear distortion model c_(NL)( ). In FIG. 2, c_(NL)( ) may be thecomposite response of DNF circuit 102, DNF PSD shaper 104, PAcompensation circuit 106, and PA 130. Thus, the circuit 212 attempts todistort the signal a(n) in a manner that estimate/predicts thedistortion that would result from a(n) passing through the DNF circuit102, the PA compensation circuit 106, and the PA 130.

The DFT circuit 214 is operable to convert the time-domain signal j(n)to frequency-domain signal J[k] (e.g., using a fast Fourier transformalgorithm).

The bin modification circuit 216 is operable to select one or morefrequency bins of the signal J[k] to be modified, and modify theselected bin(s). Modification of a bin may comprise, for example,adjusting a real and/or imaginary component (i.e., amplitude and/orphase) of that bin. Additional details of operation of the binmodification circuit 216 are described below.

The combiner 222 comprises circuitry, for example, an adder oradder-subtractor. In the example shown, combiner 222 is configured forof one complex number from another.

The PA PSD shaper circuit 108 may operate to combine its input signalwith a pre-distortion signal in order to reject out-of-band distortiondue to the PA 130. In an example implementation, the PA PSD shapercircuit 108 may implement an iterative algorithm. The PA PSD shapercircuit 108 may enhance performance in when only a relatively accuratemodel of the PA 130 is available. As discussed in further detail withreference to FIGS. 3A-3C, in some implementation of the transmitter 101,the PA PSD shaper circuit 108 may be used without use of the othercomponents of the TX nonlinear shaper 100 (i.e., without DNF circuit102, DNF PSD shaper circuit 104, and PA compensation circuit 106).

A goal of the iterative PA PSD shaper circuit 108 is to mitigateout-of-band (OOB) distortion that may be introduced by the PA 130 and/orthe DNF circuit 102 (e.g. when the DNF circuit 102 is located between PAPSD shaper 108 and PA 130, as in the example implementations describedbelow with reference to FIGS. 3A-3C). Unlike conventional DPD, the PAPSD shaper 108 does not change the desired signal, which provides avariety of advantages.

For example, where mask requirements are more difficult than EVMrequirements (thus permitting correction of out-of-band distortion atthe expense of increasing in-band distortion), as is the case for somecommunication standards such as LTE ScFDMA, keeping the desired signalintact may reduce the power of pre-distortion signal for the same levelof out-of-band rejection (since no need to correct in-band), thusallowing the PA PSD shaper 108 to operate at stronger compression than aconventional DPD system while achieving the same level of out-of-bandrejection.

Also, unlike conventional modulation, the fact that the PA PSD shaper108 keeps the desired signal intact mean that it may be used even inconjunction with high modulations requiring low EVM, when the receiveris capable of dealing with nonlinear distortion introduced by both thePA 130 and the PA PSD shaper 108.

Furthermore, when used with the DNF circuit 102, the fact that the PAPSD shaper circuit 108 keeps the desired signal intact means thatdesired/intended nonlinearity introduced by the DNF circuit 102 is notdisturbed (e.g., not inverted).

The desired signal is a frequency region, e.g. a subset of OFDM bins. Asa non-limiting example for illustration, but not limitation, the desiredsignal in a frequency division multiple access channel (e.g. uplinkOFDMA or uplink ScFDMA) may be considered to be either: (1) the entirecarrier bandwidth including guard bands; (2) the allocation of a singleuser within the multiple concurrent OFDMA/ScFDMA allocations sharing thecarrier bandwidth; or (3) some superset or subset thereof. In the firstcase, the PA PSD shaper circuit 108 may correct only distortionviolating the transmission mask (not touching in band and guard banddistortion). In the second case, the PA PSD shaper 108 also correctsdistortion that may leak into frequency bins allocated to other users.

In an example implementation, the iterative PA PSD shaper 108 maycorrect distortion in the frequency domain (e.g. the OFDM bins domain,that is, correct distortion on an OFDM-bin-by-OFDM-bin basis) bytransmitting over frequency regions, not including the desired signal(Desired OFDM bins), a pre-distortion signal generated to cancel theout-of-band distortion introduced by DNF circuit 102, the PA 130, and/orother components of the transmitter 101. When cancellation is done inthe OFDM bins domain, the PA PSD shaper circuit 108 can restore OFDMAorthogonality despite the PA compression-induced distortion. Therefore,the TX nonlinear shaper 100 can manage both mask and in-bandorthogonality requirements.

As mentioned above, the PA PSD shaper 108 may be used independent of theother components of the Tx nonlinear shaper 100. Examples of this areshown in FIGS. 3A-3C. When applied separately, the PA PSD shaper 108 mayimprove performance even in communication systems where the receiver isnot equipped to handle deep communication signal distortion (e.g., bymeasuring and modeling the nonlinear response.) In this regard:

-   -   Assuming the composite response of the analog front-end is known        (including both linear and nonlinear components), the PA PSD        shaper 108 may be used to cancel the out-of-band distortion (to        comply with applicable transmission mask) for any communication        system.    -   In some communication systems having out-of-band mask        requirement that are tougher than error vector magnitude (EVM)        requirement, power may, due to nonlinear distortion introduced        by the PA 130, need to be reduced to maintain mask compliance        while not being limited by receiver EVM. One example is LTE        uplink (UL) single carrier FDMA (ScFDMA), the same is typical        with other mobile telephony and cellular standards. In this        case, use of the PA PSD shaper circuit 108 without the DNF        circuit 102, the DNF PSD shaper circuit 104, and the PA        compensation circuit 106 may sufficiently reduce out-of-band        distortion to meet mask requirements, at the expense of somewhat        increasing EVM. In other words, the PA PSD Shaper 108 may enable        taking advantage of the EVM headroom. Moreover, in this scenario        of not being EVM limited, loss in sensitivity as a result of the        use of the PA PSD shaper 108 may be negligible, thus not        requiring an advanced receiver that is equipped (e.g., with        nonlinearity measuring and/or modeling circuitry) to compensate        for the nonlinear distortion introduced by the PA PSD Shaper        108.    -   When the PA 130 is linearized by a Digital Pre Distortion and/or        envelope tracking circuit (DPD/ET circuit 306 of FIGS. 3A-3C),        the composite response of the PA nonlinear response is        substantially a soft limiter (i.e. linear response for signals        with power below threshold, and saturation for signal with power        above threshold). In this case, the composite response of the        DPD/ET circuit 306 and PA 130 can be specified simply by        saturation power (below which DPD/ET/PA provide a linear        response), thus simplifying the configuration and operation of        the PA PSD shaper 108. (When the PA PSD Shaper 108 is used        without the DPD/ET circuit 306, the nonlinear response of the PA        130 may be tracked and provided to the PA PSD Shaper 108.)

Referring to FIG. 3A, the example transmitter 301 comprises a genericmodulator 302, an interpolator 304, the PA PSD Shaper 108, the P/S andCP circuit 124, the filtering and/or windowing circuit 126, the DNFcircuit 102, the DAC and lowpass filter circuit 128, the DPD/ET circuit306, and the PA 130. The generic modulator 302 represents anycommunication signal modulator such as, for example, an OFDM, OFDMA,single carrier, single carrier FDMA (ScFDMA) modulator.

Referring to FIG. 3B, the transmitter 351 is similar to the transmitter101 but with the generic modulator 302 replaced by a ScFDMA modulatorcomposed of the constellation mapper 114, a serial-to-parallelconversion circuit 354, an L-DFT circuit 356, and bin mapper 116. InFIG. 3B, the DFT 208 is absent, or bypassed, since x(n) is already inthe frequency domain.

Referring to FIG. 3C, the transmitter 371 is similar to the transmitter351 but with a different implementation of the PA PSD shaping circuit108. In FIG. 3C, when the PA PSD Shaper is ready for the next S[K] frombin mapper 116, the switch 220 is configured to convey S[K] to IDFT 210.IDFT 210 then transforms S[K] to a time domain representation. Duringiteration i on symbol S[K], the output of IDFT 210 is scaled, by matrixmultiplier 374, by dc_(NL) ⁻¹@S_(i-1)(n), which is the inverse of thederivative of the composite nonlinear distortion model at the pointcorresponding to the value of S_(i-1)(n) (stored in memory 218). For thefirst iteration (i.e., i=1) on a particular symbol, the signal stored inmemory 218 may be all zeros, and, assuming the c_(NL) is normalized,dc_(NL) ⁻¹ at 0=1. For any second and subsequent iteration i on the samesymbol, the value stored in the memory 218 is S_(i-1)(n). The output ofmultiplier 374 is then added to the output of memory 218 to generateS_(i)[n]. S_(i)[n] is then distorted by circuit 212 to generateJ_(i)(n). The signal J_(i)(n) is then transformed to the frequencydomain signal J_(i)[k] by DFT circuit 214. S[k] for the current symbolis then subtracted off of J_(i)[k] by combiner 222 to generate E_(i)[k].Bin modification circuit 216 then operates on E_(i)[k] to generate thepre-distortion signal D_(i)[k], which is multiplied by −1 before beingfed back via the switch 220 for iteration i+1 on the current symbol (ifi is not the last iteration). In an example implementation, thederivative of c_(NL)( ) may be calculated using the Jacobian—viewing thecomplex function C_(NL), which has complex input and complex output(C→C), as real function with two-dimensional real input andtwo-dimensional real output (R²→R²). That is, dc_(NL) ⁻¹ may becalculated as the generalized inverse of the Jacobian matrix.

In FIGS. 3A-3C, the P/S and CP circuit 124 is present only formulti-carrier modulations (OFDM, OFDMA, SCFDMA) and not for singlecarrier modulations. Similarly, the DNF circuit 102 is optional in thetransmitters of FIGS. 3A-3C.

In FIGS. 3A-3C, the DNF 102 is optional and is as described above withrespect to FIGS. 1 and 2. In transmitters 301 and 351, however, the DNFcircuit 102 is a part of the composite nonlinear response of the DNFcircuit 102, the DPD/ET circuit 306, and the PA 130. Assuming the DPD/ETcircuit 306 effectively linearizes the PA 130 (i.e. the compositeresponse of the two is a soft limiter), the composite response of theDNF 102, DPD/ET circuit 306, and PA 130 will resemble the response ofthe DNF circuit 102 for input powers below the saturation point. If theDNF circuit 102 were omitted in transmitters 301 and 351, the compositenonlinear response of the DPD/ET circuit 306 and PA 130 is a softlimiter response. Since, in this case, the PA PSD Shaper 108 precedesthe DNF 102, it handles the nonlinearity introduced both by the DNF 102and by the linearized PA response (i.e., the soft limiter response ofthe DPD/ET circuit 306 and PA 130).

Operation of the example transmitter 301 will now be described.Transmission symbols x[n] are modulated by modulator 302 and theninterpolated to a shaper oversampling rate. The resulting samples s(n)are converted to frequency by DFT circuit 208, the resulting frequencydomain signal is denoted S[k]. In an example implementation, the PA PSDshaper 108 processes S[k] in an iterative way, using a relativelyaccurate composite nonlinear distortion model c_(NL)( ). For the firstiteration on a particular symbol, the switch 220 is configured toconnect the output of the DFT 208 to the combiner 222. The signal S[k]is combined with an initial value of D[k] (output by bin modificationcircuit 216) to result in an initial value of A[k]. A[k] is then storedto memory 218. For subsequent iterations (if any) on the same symbol,the switch 220 is configured to connect the output of memory 218 to theinput of the combiner 222. That is, for second and subsequent iterationson a particular symbol, A[k] is generated from the value of A[k]generated during the previous iteration. In each iteration, A[k] isconverted to time domain signal a(n) by IDFT circuit, then circuit 212distorts the signal a(n) according to c_(NL)( ) to generate signal j(n).The distorted signal j(n) is then converted to frequency domain signalJ[k].

In an example implementation, the frequency bins of signal J[k]corresponding to desired signal (i.e., the desired frequency bins ofsignal S[k]) are then zeroed by bin modification circuit 216, resultingin the frequency distortion signal D[k]. That is, in such animplementation, the signal D[k] is zero valued in one or more frequencybins corresponding to the desired signal and non-zero valued in otherbins. These non-zero values of the signal D[k] thus correspond tointerference cancelling signals located in undesired frequency bins(i.e., in FIG. 3A, out-of-band frequency bins and/or frequency bins thatare in-band but allocated to transmitters other than 301). The frequencydistortion signal D[k] is then subtracted from A[k] of the previousiteration (output by memory 218) and the process may be repeated for oneor more iterations (e.g. resulting in 1 to 4 iterations in total). Insuch an implementation, distortion is moved (“shaped”) from undesiredfrequency bins to desired frequency bins. Thus, upon completion of thei^(th) iteration on a particular symbol, A[k] comprises the originalS[k] combined with a pre-distortion signal which is the cumulativeresult of i values of D[k].

In an example implementation, the frequency bins of signal J[k]corresponding to undesired signal (i.e., undesired frequency bins ofsignal S[k] and/or additional frequency bins not corresponding tocomponents of signal s(n)) are then zeroed by bin modification circuit216, resulting in the frequency distortion signal D[k]. That is, in suchan implementation, the signal D[k] is zero valued in one or more binscorresponding to undesired signal and non-zero valued in one or morefrequency bins corresponding to the desired signal. These non-zerovalues of the signal D[k] thus correspond to interference cancellingsignals located in desired frequency bins (i.e., in FIG. 3A, frequencybins allocated to the transmitter 301). The frequency distortion signalD[k] is then subtracted from A[k] of the previous iteration (output bymemory 218) and the process may be repeated for one or more iterations(e.g. resulting in 1 to 4 iterations in total). Thus, upon completion ofthe i^(th) iteration on a particular symbol, A[k] comprises the originalS[k] combined with a pre-distortion signal which is the cumulativeresult of i values of D[k].

After the i^(th) iteration on a particular symbol, a next symbol S[k]arrives at the PA PSD Shaper 108 and the process repeats. Thus, D[k] isupdated for each symbol and thus is periodic with a periodicitysynchronized to the symbol timing of S[k].

In the above example, the bin modification circuit 216 gives each bin ofJ[K] a weight of either 0 or 1 which results in each bin of J[K] beingeither completely subtracted from A[K], or not at all subtracted. Thebin modification circuit 216 is not, however, limited to using onlyweights of 0 and 1. Any real or complex weights may be used. Forexample, relatively lower, non-zero real weights may be used for weakercancellation of distortion on a first set of bins, and relativelyhigher, non-zero real weights may be used for stronger cancellation ofdistortion on a second set of bins. Such weightings may be chosen, forexample, based on a priori knowledge that weaker cancellation sufficeson the first set and/or that stronger cancellation is need on the secondset. Alternately, PA PSD Shaper may change the weights dynamically fromiteration to iteration based on an amount of excess distortion in a binat the end of a previous iteration. For example, if distortion of aspecific bin is lower than a determined threshold (e.g., determinedbased on the applicable standard, based on knowledge of the intendedreceiver, and/or the like) then PA PSD shaper 108 may use a low weightfor partial distortion cancellation for that bin, but if distortion isabove the threshold, the PA PSD Shaper may apply a higher weight (e.g.up to a weight of 1 to fully cancel distortion for that bin).

In an example implementation, the bin mapper circuit 116 and/or the binmodification circuit 216 may apply a complex weight to frequency bins ofsignal S[k] corresponding to pilots. Such a weighting may enable betterchannel estimation in a receiver and thus reduce EVM in the receiver. Toexplain, the transmitter may insert pilots which a receiver may use totrain its equalizer. The pilots may be, for example, scattered among thebins of an OFDM symbol or may be on all bins of certain SC-FDMA symbols(e.g., the UL Reference signal in LTE). Pilots, however, have low-PAPRand thus do not experience significant nonlinear distortion. Therefore,equalizer training based on such low-PAPR pilots does not take intoaccount the nonlinear distortion introduced by the system (including,for example, the PA PSD Shaper 108, the DNF 102, the DPD 306, and/or thePA 130). In some instances, where the receiver is capable, this may becompensated for by providing the nonlinear model (C_(NL)) to thereceiver (e.g., via a control channel) and the receiver may thus take itinto account when configuring its equalizer and/or other circuitry. Inother instances (e.g., in the case of a conventional LTE receiver),however, the receiver is not capable of receiving C_(NL) and knowing howto use it for configuring its circuitry. In these other instances, thetransmitter may intentionally distort the pilot symbols (i.e., modifythe amplitude and/or phase of the pilots) so that equalizer training inthe receiver inherently takes the multiplicative part of the distortioninto account.

The complex weight applied to bins of S[k] corresponding to pilots isdesignated Wp here and may correspond to expectancy, over the set of allpossible transmission symbols (i.e., for OFDM, all possible OFDM symbolsor, for SC-FDMA, all possible SC-FDMA symbols), of the amplitudereduction and phase rotation caused by the nonlinear elements of thetransmitter. The expectancy may be calculated from C_(NL) using a model,formula, or lookup table, for example. The expectancy may depend on theparticular allocation of bins among the users.

Although the complex weighting of pilots may occur in the frequencydomain, as just described, in other implementations it may be applied inthe time domain. Complex weighting of pilots in the time domain may beparticularly straightforward where all bins of a particular symbol carrypilots, such as in an LTE UL Reference Signal. This is shown in FIG. 3Dwhere multiplier 390 multiplies S(n) by 1 for data symbols and by W_(p)for pilot symbols.

In an example implementation (e.g., where the transmitter is in abasestation using downlink (DL) OFDMA), the distortion may be shaped bymoving it to frequencies allocated for use with relatively lower-ordermodulation constellations. For example, if out-of-band distortion is toohigh and is desired to be moved in-band, the frequencies onto which tomove that distortion may be selected to be frequencies allocated for usewith relatively lower-order modulation constellations. Similarly, ifdistortion in a first frequency bin is too high and it is desired tomove it to a second one or more bin(s), those second bins may beselected from subcarriers using a lower-order modulation constellationthan the constellation being used on the first bin. (e.g., thehigher-order constellation may be NQAM and the lower order constellationmay be MQAM with N>M). Such distortion shaping may be selected to takeadvantage of the fact that frequencies allocated for use withhigher-order modulation constellations would typically require lowerdistortion floors than the frequencies allocated for use with thelower-order modulation constellations. Such shaping may be achieved bymore aggressive cancelling of distortion on frequencies allocated foruse with higher-order modulation constellation allocations and lessaggressive cancelling (or no cancelling at all) on frequencies allocatedfor use with lower-order modulation constellations. The PA PSD Shaper108 may also operate to move distortion from in-band to out-of-band incases where very low in-band EVM is needed, and out-of-band rejectionrequirements can tolerate some additional out-of-band distortion.

In an example implementation (e.g., where the transmitter is in ahandset using uplink (UL) ScFDMA), different weights to be applied todifferent in-band allocated frequencies may be determined based on knownnoise and/or interference at the receiver. For example, if the noiseplus interference floor at receiver is not flat by design of the network(e.g., in a network using fractional frequency re-use), such that thereis lower interference on some frequencies than others, the binmodification circuit 216 may take this into account when determiningweights to apply to the various frequencies.

Operation of the example transmitter 351 will now be described. A bitstream b[ ] is constellation mapped (e.g., according to a selected QAMconstellation) into the symbol stream x[n]. S/P circuit 354 groups thesymbol stream into groups of size L, where L is the number of frequencybins allocated for ScFDMA burst transmission. A L-size DFT is performedby L-DFT circuit 356 to transform ScFDMA symbols into L frequency domainbins. The bin mapper 116 then maps the resulting L bins according tospecific burst allocation in frequency (i.e., maps the bins to thefrequencies allocated to the transmitter 351). The output of the binmapper 116 represents a DPD oversampled signal, and consists ofNfft*DPD_OVS frequency bins (where Nfft corresponds to ScFDMA carrierBW, and DPD_OVS is the DPD oversampling rate). This frequency domainsignal is taken directly as signal S[k] and input to the PA PSD shaper108. The rest of the processing blocks are the same as described forFIG. 3A. This results in the pre-distortion cancellation signal(accumulation of D[k]) being non-overlapping in frequency andsynchronized in time to the desired ScFDMA transmission, andtherefore—disregarding non-linearity—orthogonal at the base stationreceiver to the desired ScFDMA transmission. This type of cyclical andsynchronized pre-distortion ensures ScFDMA orthogonality despitenonlinear distortion introduced in the transmitter.

In an example implementation, D[k] for the first iteration may beconfigured based on a priori knowledge about the communication system.In another example implementation, D[k] for the first iteration may besimply all zeroes. In such an implementation, the IDFT circuit 210 maybe skipped for the first iteration and circuit 212 may operate directlyon s(n) for the first iteration.

In the example implementations of FIGS. 3A-3C, the PA PSD Shaper 108 isfollowed by adding CP, filtering and OFDM windowing, optionally applyingthe Digital Nonlinear Function (DNF), and then digital-to-analogconversion. The DAC output is optionally filtered and processed byDPD/ET which drives the input of the PA 130. The DNF circuit 102 andDPD/ET circuit 306 are optional, and thus any combination of DNF circuit102, DPD/ET circuit 306, and PA 130 determines the composite nonlinearresponse c_(NL)( ). That is, the composite nonlinear distortion modelc_(NL)( ) may, for example, be a model of the PA 130 (i.e., c_(NL)()=ƒ_(NL)( )) if neither DPD/ET circuit 306 nor DNF circuit 102 are inuse, a soft limiter model when using DPD/ET circuit 306 in combinationwith the PA 130, or the response of the DNF circuit 102 when using theDNF 102, DPD/ET circuit 306, and PA 130.

When the DPD/ET circuit 306 is present and DNF circuit 102 is not used(i.e., not present or bypassed), then c_(NL)( )=ƒ_(NL)( ) (an estimationof the nonlinear distortion introduced by PA 130). This model may be amemoryless nonlinearity (in the simpler case), or consist ofnonlinearity with memory. Estimation of ƒ_(NL)( ) may, for example, bebased on feedback from the PA 130 and/or based on a priori informationabout the PA 130 (e.g., its part number, its bias voltage,characterization data from a manufacturer of the PA, etc.).

When the DPD/ET circuit 306 is present, it may use feedback from PA 130to estimate, and compensate for, ƒ_(NL)( ). This may result in acomposite response that is approximately a soft-limiter (an examplesoft-Limiter response is shown in FIG. 4). In other words, in this casethe c_(NL)( ) used by the PA PSD Shaper 108 may be either a soft limiteror approximately a soft limiter. In the case that the composite responseis approximately a soft limiter, the response of the PA may not becompletely inverted and the result may be a time invariant response thatmay compress when the signal approaches saturation. The compressioneffect may or may not have memory (i.e., depend on previous data). Inthe case that the composite response is approximately a soft limiter, arepresentation of the composite nonlinearity may be conveyed from theDPD/ET circuit 306 to the PA PSD shaper 108. In the case that thecomposite response is a soft-limiter, only power amplifier input powerbackoff may need to be conveyed to the PA PSD Shaper 108, thus having avery simple interface. In some implementation, however, a soft limiterresponse (or approximation thereof), may not provide the bestperformance when used in conjunction with the PA PSD Shaper 108. In suchimplementations, the DNF circuit 102 may be used to turn the softlimiter response (or approximation thereof) to a response that providesbetter performance in conjunction with the PA PSD Shaper 108. In thiscase, the composite non-linearity c_(NL)( ) is composed of the responseof the DNF circuit 102, the response of the DPD/ET circuit 306, and theresponse of the PA 130, and may be computed by the DNF circuit 102processing the composite nonlinearity model provided by the DPD/ETcircuit 306.

In an example implementation, the PA PSD shaper 108 may use numericaloptimization to, for example, determine which weights to apply tovarious bins of J[k]. This may comprise optimization of a performancemetric using numerical techniques such as gradient descent. One exampleof such a performance metric is the mean squared error betweenpermissible out-of-band distortion (as set by an applicable standardand/or regulatory body) and the out-of-band distortion predicted byD[k]. The result of the numerical optimization may be that the binmodification circuit 216 weights differently distortion in differentfrequency bands. For example, the bin modification circuit 216 may applyhigher weights to out-of-band distortion components exceeding anapplicable spectral mask and lower weights to distortion componentsleaking into bins allocated to other users. The use of numericaloptimization may use, for example, relatively few gradient descentiterations, thus resulting in a cost-effective implementation.

It is noted that, in the example implementations of FIGS. 1-3C, the TXnonlinear shaping operates cyclically (i.e. CP extended), in order tomaintain orthogonality to the desired OFDM signal and possibly weightdistortion of different OFDM bins differently. However, in anotherimplementation, the TX nonlinear shaping may be applied after adding thecyclic prefix. Such an implementation is also applicable when CP isreplaced by PRBS as in TDS-OFDM, but, in that case, the resulting PRBSinterference into the desired signal may need to be corrected (as is thecase with conventional PRBS CP systems).

FIG. 5 is a diagram illustrating control of the PA PSD Shaper inaccordance with an example implementation of this disclosure. Shown inFIG. 5 is a block 502 representing circuitry operable to configure andcontrol operation of the PA PSD Shaper 108. The control circuitry 502may, for example, be implemented by a microcontroller of thetransmitter, by dedicated control circuitry, and/or by the PA PSD Shaper108 itself.

The control circuitry 502 is operable to configure the switch 220. Thiscontrol circuit is also operable to control when the circuit 124 readsin the output of the PA PSD Shaper 108. Through these two controls, thecontrol circuitry 502 can thus control the number of iterationsperformed on any given symbol. The number of iterations performed on anyparticular symbol may be determined a priori and/or in real-time. Thenumber of iterations performed on any particular symbol may bedetermined based on any one or more of a variety of parameters such as:the constellation used to generate the symbol, the particularconstellation points in the symbol, the peak to average power ratio(PAPR) of the symbol, the number of subcarriers in the symbol, whichsubset of a larger subset of subcarriers have been allocated fortransmission of the symbol, a spectral mask of the standard with whichtransmission of the symbol is to comply, a permissible error vectormagnitude (e.g., set by an applicable standard and/or by thecapabilities of the receiver for which the symbol is destined), thepower backoff setting of the power amplifier, a performance metric,and/or the like. A performance metric used to control the number ofiterations may be measured by the transmitter and/or may be measured bya receiver and fed back to the transmitter via a control channel. Aperformance metric used to control the number of iterations may be, forexample, an error (e.g., a difference or squared difference) between thesignal S(n) or J(n) and a requirement set forth by an applicablestandard (e.g., maximum out-of-band power, EVM, and/or the like).

The control circuitry is also operable to configure the bin modificationcircuit 216. This may comprise selecting which of the bins may bemodified and the extent to which the selected bins may be modified.Configuration of the bin modification circuit 216 may be determined apriori and/or in real-time (e.g., on a per-symbol or per-link basis).The configuration of the bin modification circuit 216 for any particularsymbol may be determined based on any one or more of a variety ofparameters such as: the constellation used to generate the symbol, theparticular constellation points in the symbol, the peak to average powerratio (PAPR) of the symbol, the number of subcarriers in the symbol,which subset of a larger subset of subcarriers have been allocated fortransmission of the symbol, a spectral mask of the standard with whichtransmission of the symbol is to comply, a permissible error vectormagnitude (e.g., set by an applicable standard and/or by thecapabilities of the receiver for which the symbol is destined), thepower backoff setting of the power amplifier, a performance metric,and/or the like. A performance metric used to for configuring the binmodification circuit 216 (e.g., configuring weights applied to variousbins for any particular iteration i on a symbol) may be measured by thetransmitter and/or may be measured by a receiver and fed back to thetransmitter via a control channel. A performance metric used toconfigure the bin modification circuit 216 may be, for example, an error(e.g., a difference or squared difference) between the signal S(n) orJ(n) and a requirement set forth by an applicable standard (e.g.,maximum out-of-band power, EVM, and/or the like).

In an example implementation, processing

FIG. 6A illustrates an alternate implementation of the transmitter ofFIG. 3A. Relative to the implementation shown in FIG. 3A, theimplementation in FIG. 6B additionally comprises IDFT 602, accumulator604, and combiner 606. The IDFT 602 converts D[k] to a time domainrepresentation D(n). The accumulator 604 then sums the values of D[k]over the iterations on symbol s(n). After the last iteration on symbols(n), the pre-distortion signal output by the accumulator 604 issubtracted from s(n) by combiner 606. In yet another implementation,D[k] may be accumulated in the frequency domain prior to conversion totime-domain representation which may then be subtracted from s(n).

FIG. 6B illustrates an alternate implementation of the transmitter ofFIG. 3B. Relative to the implementation shown in FIG. 3B, theimplementation in FIG. 6B additionally comprises IDFT 602, accumulator604, combiner 606, and switch 608. The IDFT 602 converts D[k] to a timedomain representation D(n). The switch 608 routes D(n) to accumulator604 which sums the values of D[k] over the iterations on symbol s(n).After the last iteration on symbol s(n), switch 608 routes s(n) to thecombiner 606 and the pre-distortion signal output by the accumulator 604is subtracted from s(n) by combiner 606. In yet another implementation,D[k] may be accumulated in the frequency domain prior to conversion totime-domain representation which may then be subtracted from s(n).

In accordance with an example implementation of this disclosure, atransmitter (e.g., 301 or 351) comprises at least one nonlinear circuit,and a power spectral density (PSD) shaping circuit (e.g., 108). The PSDshaping circuit is operable to receive a symbol of a modulated signal(e.g., time domain symbol s(n) or frequency domain symbol s[k]), whereinthe symbol corresponds to a first one or more frequency bins. The PSDshaping circuit is operable to perform iterative processing of thesymbol, wherein each iteration of the processing comprises: generationof a first pre-distortion signal (e.g., frequency domain signal D[k] ortime-domain signal D(n)) based on a model of the at least one nonlinearcircuit, wherein the pre-distortion signal corresponds to a second oneor more frequency bins; and combination of the symbol (e.g., Sap, or apre-distorted version of the symbol (e.g., A[k]), with thepre-distortion signal. The generation of the pre-distortion signal maycomprises generation of a nonlinearly-distorted signal (e.g., J(n) orJ[k]), and adjustment of one or more components of thenonlinearly-distorted signal.

The adjustment may comprise weighting the one or more components of thenonlinearly-distorted signal. The weighting may be determined perfrequency bin or per group of frequency bins. The weighting may bedetermined based on distortion remaining in the pre-distorted signalafter an iteration of the iterative processing of the symbol. The one ormore components may be in one or more desired signal frequency bands.The one or more desired signal frequency bands may be frequency bandsassigned to the transmitter by a basestation. The one or more componentsmay be in one or more undesired signal frequency bands. The one or moreundesired signal frequency bands may be out-of-band frequency bands. Theone or more undesired signal frequency bands may be frequency bandsallocated, by a basestation, to one or more other transmitters. Theadjustment of the one or more components may be controlled (e.g., bycircuit 502) based on a performance metric (e.g., SNR) fed back to thetransmitter from a receiver. The adjustment of the one or morecomponents may be controlled based on order of a modulationconstellation used for generation of the symbol. The adjustment of theone or more components may be controlled based on a determination ofinterference present at a receiver to which said symbol is to betransmitted.

The adjustment of the one or more components may be controlled based onwhich frequency bands have been allocated to the transmitter by abasestation. The adjustment of the one or more components may results inshifting distortion from out-of-band frequencies to in-band frequenciesor from in-band frequencies to out-of-band frequencies. The adjustmentof the one or more components may result in shifting distortion fromin-band frequencies to out-of-band frequencies. The adjustment of theone or more components may be controlled based on whether a receiver towhich the symbol is destined comprises circuitry for modeling andmitigating nonlinear distortion introduced by the transmitter. The atleast one nonlinear circuit may comprise a power amplifier and/or adigital pre-distortion circuit. The at least one nonlinear circuit maycomprise a digital nonlinear function circuit configured to compress asignal to be transmitted prior to input of the signal to be transmittedto a power amplifier of the transmitter. The said generation of thepre-distortion signal may be based on a seeking of a minimum, maximum,or desired value (e.g., above or below a threshold) of a performancemetric. The transmitter may comprise a pilot distortion circuit (e.g.,bin mapper 116 or multiplier 390) operable to distort pilots of saidsymbol.

In accordance with an example implementation of this disclosure, atransmitter (e.g., 301 or 351) may comprise a modulator circuit, acombiner circuit (e.g., 222 and/or 606), a nonlinear distortion modelingcircuit (e.g., 212), and a modification circuit (e.g., 216). Themodulator is operable to generate a symbol to be transmitted (e.g., timedomain symbol s(n) or frequency domain symbol sap. The combiner circuitis operable to combine the symbol with a first portion of apre-distortion signal (e.g., D[k] for a first iteration), a result ofthe combination being a pre-distorted signal (e.g., a(n)). The nonlineardistortion modeling circuit is operable to: model nonlinear distortionintroduced by at least one nonlinear circuit of the transmitter; andnonlinearly distort the pre-distorted signal, a result of the nonlineardistortion being a distorted signal (e.g., J(n) or Jai). Themodification circuit is operable to modify the distorted signal togenerate a second portion of a pre-distortion signal (e.g., D[k] for asecond iteration). The modification of the distorted signal may be on aper subcarrier basis. The modification of the distorted signal by themodification circuit may comprise zeroing spectral components of thedistorted signal.

As utilized herein the terms “circuits” and “circuitry” refer tophysical electronic components (i.e. hardware) and any software and/orfirmware (“code”) which may configure the hardware, be executed by thehardware, and or otherwise be associated with the hardware. As usedherein, for example, a particular processor and memory may comprise afirst “circuit” when executing a first one or more lines of code and maycomprise a second “circuit” when executing a second one or more lines ofcode. As utilized herein, “and/or” means any one or more of the items inthe list joined by “and/or”. As an example, “x and/or y” means anyelement of the three-element set {(x), (y), (x, y)}. In other words, “xand/or y” means “one or both of x and y”. As another example, “x, y,and/or z” means any element of the seven-element set {(x), (y), (z), (x,y), (x, z), (y, z), (x, y, z)}. In other words, “x, y and/or z” means“one or more of x, y and z”. As utilized herein, the term “exemplary”means serving as a non-limiting example, instance, or illustration. Asutilized herein, the terms “e.g.,” and “for example” set off lists ofone or more non-limiting examples, instances, or illustrations. Asutilized herein, circuitry is “operable” to perform a function wheneverthe circuitry comprises the necessary hardware and code (if any isnecessary) to perform the function, regardless of whether performance ofthe function is disabled or not enabled (e.g., by a user-configurablesetting, factory trim, etc.).

Other embodiments of the invention may provide a non-transitory computerreadable medium and/or storage medium, and/or a non-transitory machinereadable medium and/or storage medium, having stored thereon, a machinecode and/or a computer program having at least one code sectionexecutable by a machine and/or a computer, thereby causing the machineand/or computer to perform the processes as described herein.

Accordingly, the present invention may be realized in hardware,software, or a combination of hardware and software. The presentinvention may be realized in a centralized fashion in at least onecomputing system, or in a distributed fashion where different elementsare spread across several interconnected computing systems. Any kind ofcomputing system or other apparatus adapted for carrying out the methodsdescribed herein is suited. A typical combination of hardware andsoftware may be a general-purpose computing system with a program orother code that, when being loaded and executed, controls the computingsystem such that it carries out the methods described herein. Anothertypical implementation may comprise an application specific integratedcircuit or chip.

The present invention may also be embedded in a computer programproduct, which comprises all the features enabling the implementation ofthe methods described herein, and which when loaded in a computer systemis able to carry out these methods. Computer program in the presentcontext means any expression, in any language, code or notation, of aset of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiment disclosed, but that the present invention willinclude all embodiments falling within the scope of the appended claims.

1-20. (canceled)
 21. A system comprising: combiner circuitry configuredto combine a first signal and a second signal to generate a thirdsignal, wherein said third signal is output for transmission onto acommunication medium; nonlinearity modeling circuitry configured to:model a nonlinear response of a nonlinear circuit; and generate a fourthsignal through nonlinear distortion of a fifth signal using said model;and bin modification circuitry configured to: generate said secondsignal through application of one or more weighting factors to frequencybins of said fourth signal, or of a transformed version of said fourthsignal; and determine said weighting factors based on a difference, orsquared difference, error metric.
 22. The system of claim 21, whereinsaid fifth signal is a transformed version of said third signal.
 23. Thesystem of claim 21, wherein: said one or more weighting factorscomprises a plurality of weighting factors; a first weighting factor ofsaid plurality of weighting factors is different than a second weightingfactor of said plurality of weighting factors; and said first weightingfactor is applied to a first frequency bin of said fourth signal andsaid second weighting factor is applied to a second frequency bin ofsaid fourth signal.
 24. The system of claim 23, wherein said firstfrequency bin is a desired frequency bin and said second frequency binis an undesired frequency bin.
 25. The system of claim 23, wherein firstfrequency bin is an in-band frequency bin and said second frequency binis an out-of-band frequency bin.
 26. The system of claim 21, whereinsaid combiner circuitry, said nonlinearity modeling circuitry, and saidbin modification circuitry reside in a transmitter.
 27. The system ofclaim 26, wherein: said one or more weighting factors comprises aplurality of weighting factors; a first weighting factor of saidplurality of weighting factors is different than a second weightingfactor of said plurality of weighting factors; said first weightingfactor is applied to a first frequency bin of said fourth signal andsaid second weighting factor is applied to a second frequency bin ofsaid fourth signal; said first frequency bin is assigned to saidtransmitter by a basestation; and said second frequency bin is assignedto another transmitter by said basestation.
 28. The system of claim 21,wherein said bin modification circuitry is configured to adjust said oneor more weighting factors based on a performance metric fed back to saidtransmitter from a receiver.
 29. The system of claim 21, wherein saidbin modification circuitry is configured to adjust said one or moreweighting factors based on order of a modulation constellation used forgeneration of said symbol.
 30. The system of claim 21, wherein said binmodification circuitry is configured to adjust said one or moreweighting factors based on a determination of interference present at areceiver to which said symbol is to be transmitted.
 31. A methodcomprising: combining, by combiner circuitry of a transmitter, a firstsignal and a second signal to generate a third signal, wherein saidthird signal is output for transmission onto a communication medium;generating, by nonlinearity modeling circuitry of said transmitter, amodel of a nonlinear response of a nonlinear circuit of saidtransmitter; generating, by said nonlinearity modeling circuitry, afourth signal through nonlinear distortion of a fifth signal using saidmodel; generating, by bin modification circuitry of said transmitter,said second signal through application of one or more weighting factorsto frequency bins of said fourth signal, or of a transformed version ofsaid fourth signal; and determining, by said bin modification circuitry,said weighting factors based on a difference, or squared difference,error metric.
 32. The method of claim 31, wherein said fifth signal is atransformed version of said third signal.
 33. The method of claim 31,wherein: said one or more weighting factors comprises a plurality ofweighting factors; a first weighting factor of said plurality ofweighting factors is different than a second weighting factor of saidplurality of weighting factors; and said first weighting factor isapplied to a first frequency bin of said fourth signal and said secondweighting factor is applied to a second frequency bin of said fourthsignal.
 34. The method of claim 33, wherein said first frequency bin isa desired frequency bin and said second frequency bin is an undesiredfrequency bin.
 35. The method of claim 33, wherein first frequency binis an in-band frequency bin and said second frequency bin is anout-of-band frequency bin.
 36. The method of claim 31, wherein saidcombiner circuitry, said nonlinearity modeling circuitry, and said binmodification circuitry reside in a transmitter.
 37. The method of claim36, wherein: said one or more weighting factors comprises a plurality ofweighting factors; a first weighting factor of said plurality ofweighting factors is different than a second weighting factor of saidplurality of weighting factors; said first weighting factor is appliedto a first frequency bin of said fourth signal and said second weightingfactor is applied to a second frequency bin of said fourth signal; saidfirst frequency bin is assigned to said transmitter by a basestation;and said second frequency bin is assigned to another transmitter by saidbasestation.
 38. The method of claim 31, comprising adjusting, by saidbin modification circuitry, said one or more weighting factors based ona performance metric fed back to said transmitter from a receiver. 39.The method of claim 31, comprising adjusting, by said bin modificationcircuitry, said one or more weighting factors based on order of amodulation constellation used for generation of said symbol.
 40. Themethod of claim 21, comprising adjusting, by said bin modificationcircuitry, said one or more weighting factors based on a determinationof interference present at a receiver to which said symbol is to betransmitted.